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LTC3558 Linear USB Battery Charger with Buck and Buck-Boost Regulators FEATURES
Battery Charger n Standalone USB Charger n Up to 950mA Charge Current Programmable via Single Resistor n HPWR Input Selects 20% or 100% of Programmed Charge Current n NTC Input for Temperature Qualified Charging n Internal Timer Termination n Bad Battery Detection Switching Regulators (Buck and Buck-Boost) n Up to 400mA Output Current per Regulator n 2.25MHz Constant-Frequency Operation n Power Saving Burst Mode(R) Operation n Low Profile, 20-Lead, 3mm x 3mm QFN Package
DESCRIPTION
The LTC(R)3558 is a USB battery charger with dual high efficiency switching regulators. The device is ideally suited to power single-cell Li-Ion/Polymer based handheld applications needing multiple supply rails. Battery charge current is programmed via the PROG pin and the HPWR pin with capability up to 950mA of current at the BAT pin. The CHRG pin allows battery status to be monitored continuously during the charging process. An internal timer controls charger termination. The part includes monolithic synchronous buck and buckboost regulators that can provide up to 400mA of output current each and operate at efficiencies greater than 90% over the entire Li-Ion/Polymer battery range. The buckboost regulator can regulate its programmed output voltage at its rated deliverable current over the entire Li-Ion range without drop out, increasing battery runtime. The LTC3558 is offered in a low profile (0.75mm), thermally enhanced, 20-lead (3mm x 3mm) QFN package.
, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.
APPLICATIONS
n n
SD/Flash-Based MP3 Players Low Power Handheld Applications
TYPICAL APPLICATION
USB Charger Plus Buck Regulator and Buck-Boost Regulator
USB (4.3V TO 5.5V) 1F 1.74k PROG NTC LTC3558 CHRG SUSP DIGITAL CONTROL HPWR MODE EN1 EN2 FB2 105k EXPOSED PAD 15k 330pF VC2
3558 TA01
Demo Board
VCC
BAT PVIN1 PVIN2 10F
+
SINGLE Li-lon CELL (2.7V TO 4.2V) 1.2V AT 400mA
SW1 FB1 SWAB2
4.7H
324k 649k
10pF
10F
2.2H 3.3V AT 400mA SWCD2 VOUT2 324k 121k 33pF 22F
10pF
GND
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LTC3558 ABSOLUTE MAXIMUM RATINGS
(Note 1)
PIN CONFIGURATION
TOP VIEW HPWR 15 EN2 21 14 VC2 13 FB2 12 SUSP 11 VOUT2 6 SW1 7 PVIN1 8 PVIN2 9 10 SWAB2 SWCD2 CHRG PROG NTC VCC GND 1 BAT 2 MODE 3 FB1 4 EN1 5
VCC (Transient); t < 1ms and Duty Cycle < 1%....................... -0.3V to 7V VCC (Static) .................................................. -0.3V to 6V BAT, CHRG ................................................... -0.3V to 6V PROG, SUSP .................................-0.3V to (VCC + 0.3V) HPWR, NTC................... -0.3V to Max (VCC, BAT) + 0.3V PROG Pin Current ...............................................1.25mA BAT Pin Current ..........................................................1A PVIN1, PVIN2 ..................................-0.3V to (BAT + 0.3V) EN1, EN2, MODE, VOUT2 .............................. -0.3V to 6V FB1, SW1 ......................... -0.3V to (PVIN1 + 0.3V) or 6V FB2, VC2, SWAB2 ............. -0.3V to (PVIN2 + 0.3V) or 6V SWCD2 ............................-0.3V to (VOUT2 + 0.3V) or 6V ISW1 ...............................................................600mA DC ISWAB2, ISWCD2, IVOUT2 ...................................750mA DC Junction Temperature (Note 2) ............................. 125C Operating Temperature Range (Note 3).... -40C to 85C Storage Temperature.............................. -65C to 125C
20 19 18 17 16
UD PACKAGE 20-LEAD (3mm x 3mm) PLASTIC QFN TJMAX = 125C, JA = 68C/W EXPOSED PAD (PIN 21) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LTC3558EUD#PBF TAPE AND REEL LTC3558EUD#TRPBF PART MARKING LDCD PACKAGE DESCRIPTION 20-Lead (3mm x 3mm) Plastic QFN TEMPERATURE RANGE -40C to 85C Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
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LTC3558 ELECTRICAL CHARACTERISTICS
SYMBOL VCC IVCC VFLOAT ICHG IBAT PARAMETER Input Supply Voltage Battery Charger Quiescent Current (Note 4) BAT Regulated Output Voltage Constant-Current Mode Charge Current Battery Drain Current Standby Mode, Charge Terminated Suspend Mode, VSUSP = 5V 0C TA 85C l HPWR = 1 HPWR = 0 Standby Mode, Charger Terminated, EN1 = EN2 = 0 Shutdown, VCC < VUVLO, BAT = 4.2V, EN1 = EN2 = 0 Suspend Mode, SUSP = 5V, BAT = 4.2V, EN1 = EN2 = 0 VCC = 0V, EN1 = EN2 = 1, MODE = 1, FB1 = FB2 = 0.85V, VOUT2 = 3.6V BAT = 3.5V, VCC Rising BAT = 3.5V BAT = 4.05V, (VCC - BAT) Falling BAT = 4.05V HPWR = 1 HPWR = 0 BAT < VTRKL BAT < VTRKL BAT Rising Threshold Voltage Relative to VFLOAT BAT Falling BAT = VFLOAT BAT < VTRKL IBAT Falling IBAT = 190mA 3.5 0.4 0.085 36 2.8 -75 30 4.179 4.165 440 84 Battery Charger
l
The l denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25C. VCC = 5V, BAT = PVIN1 = PVIN2 = 3.6V, RPROG = 1.74k, unless otherwise noted.
CONDITIONS MIN 4.3 285 8.5 4.200 4.200 460 92 -3.5 -2.5 -1.5 -50 4 200 50 130 1.000 0.200 0.100 800 46 2.9 100 -95 1.7 4 0.5 0.1 2.2 500 105 4.5 0.6 0.11 -115 56 3 70 TYP MAX 5.5 400 17 4.221 4.235 500 100 -7 -4 -3 -100 4.125 UNITS V A A V V mA mA A A A A V mV mV mV V V V mA/mA mA V mV mV ms Hour Hour mA/mA ms m C
VUVLO VUVLO VDUVLO VDUVLO VPROG hPROG ITRKL VTRKL VTRKL VRECHRG tRECHRG tTERM tBADBAT hC/10 tC/10 RON(CHG) TLIM NTC VCOLD VHOT VDIS INTC
Undervoltage Lockout Threshold Undervoltage Lockout Hysteresis Differential Undervoltage Lockout Threshold Differential Undervoltage Lockout Hysteresis PROG Pin Servo Voltage
3.85
Ratio of IBAT to PROG Pin Current Trickle Charge Current Trickle Charge Threshold Voltage Trickle Charge Hysteresis Voltage Recharge Battery Threshold Voltage Recharge Comparator Filter Time Safety Timer Termination Period Bad Battery Termination Time End-of-Charge Comparator Filter Time Battery Charger Power FET OnResistance (Between VCC and BAT) Junction Temperature in Constant Temperature Mode Cold Temperature Fault Threshold Voltage Hot Temperature Fault Threshold Voltage NTC Disable Threshold Voltage NTC Leakage Current
End-of-Charge Indication Current Ratio (Note 5)
Rising NTC Voltage Hysteresis Falling NTC Voltage Hysteresis Falling NTC Voltage Hysteresis VNTC = VCC = 5V
75 33.4
l
0.7 -1
76.5 1.6 34.9 1.6 1.7 50
78 36.4 2.7 1
%VCC %VCC %VCC %VCC %VCC mV A
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LTC3558 ELECTRICAL CHARACTERISTICS
SYMBOL VIL VIH RDN VCHRG ICHRG PVIN1 IPVIN1 PARAMETER Input Low Voltage Input High Voltage Logic Pin Pull-Down Resistance CHRG Pin Output Low Voltage CHRG Pin Input Current Input Supply Voltage Pulse Skip Input Current Burst Mode Current Shutdown Current Supply Current in UVLO PVIN1 Falling PVIN1 Rising Switching Frequency Peak PMOS Current Limit Feedback Voltage FB Input Current Maximum Duty Cycle RDS(ON) of PMOS RDS(ON) of NMOS SW Pull-Down in Shutdown Input Supply Voltage PWM Input Current Burst Mode Input Current Shutdown Current Supply Current in UVLO PVIN2 Falling PVIN2 Rising Minimum Regulated Buck-Boost VOUT Maximum Regulated Buck-Boost VOUT Forward Current Limit (Switch A) Forward Current Limit (Switch A) Reverse Current Limit (Switch D) Maximum Deliverable Output Current in Burst Mode Operation Feedback Servo Voltage FB2 Input Current Switching Frequency MODE = 0 MODE = 1 MODE = 0 MODE = 1 2.7V < PVIN2 < 4.2V 2.75V < VOUT2 < 5.5V
l l l l l l
The l denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25C. VCC = 5V, BAT = PVIN1 = PVIN2 = 3.6V, RPROG = 1.74k, unless otherwise noted.
CONDITIONS HPWR, SUSP MODE, EN1, EN2 Pins , HPWR, SUSP MODE, EN1, EN2 Pins , HPWR, SUSP Pins ICHRG = 5mA BAT = 4.5V, VCHRG = 5V
l l
MIN
TYP
MAX 0.4
UNITS V V M mV A V A A A A V V MHz mA mV nA % k
Logic (HPWR, SUSP CHRG, EN1, EN2, MODE) , 1.2 1.9 4 100 0 2.7 220 35 0 4 2.45 2.55 2.25 800 800 6.3 250 1 4.2 400 50 2 8 2.70 2.59 1050 820 50 0.65 0.75 13 2.7 220 20 0 4 2.45 2.55 2.65 5.60 700 250 450 0 820 320 575 35 4.2 400 30 1 8 2.70 2.75
Buck Switching Regulator FB1 = 0.85V, MODE = 0 (Note 6) FB1 = 0.85V, MODE = 1 (Note 6) EN1 = 0 PVIN1 = PVIN2 = 2V
l l l
PVIN1 UVLO fOSC ILIMSW1 VFB1 IFB1 DMAX1 RPMOS1 RNMOS1 RSW1(PD) PVIN2 IPVIN2
2.30 1.91 550
MODE = 0 MODE = 0 FB1 = 0.85V FB1 = 0V ISW1 = 100mA ISW1 = -100mA
l l
780 -50 100
Buck-Boost Switching Regulator V A A A A V V V V mA mA mA mA mA 800 2.25 820 50 2.59 mV nA MHz MODE = 0, IOUT = 0A, FB2 = 0.85V (Note 6) MODE = 1, IOUT = 0A, FB2 = 0.85V (Note 6) EN2 = 0, IOUT = 0A PVIN1 = PVIN2 = 2V
l l
PVIN2 UVLO VOUT2(LOW) VOUT2(HIGH) ILIMF2 IPEAK2(BURST) ILIMR2 IMAX2(BURST) VFB2 IFB2 fOSC
2.30
5.45 580 180 325 -35 50 780 -50 1.91
IZERO2(BURST) Reverse Current Limit (Switch D)
FB2 = 0.85V MODE = 0
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LTC3558 ELECTRICAL CHARACTERISTICS
SYMBOL RDSP(ON) RDSN(ON) ILEAK(P) ILEAK(N) DCBUCK(MAX) tSS2 ROUT(PD) PARAMETER PMOS RDS(ON) NMOS RDS(ON) PMOS Switch Leakage NMOS Switch Leakage Maximum Buck Duty Cycle Soft-Start Time VOUT Pull-Down in Shutdown Switches A, D Switches B, C MODE = 0 MODE = 0
l
The l denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25C. VCC = 5V, BAT = PVIN1 = PVIN2 = 3.6V, RPROG = 1.74k, unless otherwise noted.
CONDITIONS VOUT = 3.6V -1 -1 100 75 0.5 10 MIN TYP 0.6 0.6 1 1 MAX UNITS A A % % ms k
DCBOOST(MAX) Maximum Boost Duty Cycle
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD * JA) Note 3: The LTC3558E is guaranteed to meet specifications from 0C to 85C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls.
Note 4: VCC supply current does not include current through the PROG pin or any current delivered to the BAT pin. Total input current is equal to this specification plus 1.00125 * IBAT where IBAT is the charge current. Note 5: IC/10 is expressed as a fraction of measured full charge current with indicated PROG resistor. Note 6: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency.
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LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS
Suspend State Supply and BAT Currents vs Temperature
10 9 8 7 CURRENT (A) VFLOAT (V) 6 5 4 3 2 1 0 -55 -35 25 5 -15 45 TEMPERATURE (C) 65 85 IBAT 4.18 4.17 4.16 -55 -35 -15 45 25 5 TEMPERATURE (C) 65 85
3558 G02
TA = 25C, unless otherwise noted. Battery Regulation (Float) Voltage vs Battery Charge Current, Constant-Voltage Charging
4.205 4.200 4.195 4.190 4.185 VBAT (V) 4.180 4.175 4.170 4.165 4.160 4.155 4.150 VCC = 5V HPWR = 5V RPROG = 845 EN1 = EN2 = 0V 0 100 200 300 400 500 600 700 800 900 1000 IBAT (mA)
3558 G03
Battery Regulation (Float) Voltage vs Temperature
4.24 VCC = 5V 4.23 4.22 4.21 4.20 4.19
IVCC
VCC = 5V BAT = 4.2V SUSP = 5V EN1 = EN2 = 0V
3558 G01
Battery Charge Current vs Supply Voltage
500 VCC = 5V 495 HPWR = 5V 490 RPROG = 1.74k 485 EN1 = EN2 = 0V 480 IBAT (mA) 470 465 460 455 450 445 440 4.3 4.4 4.5 4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 VCC (V)
3558 G04
Battery Charge Current vs Battery Voltage
500 450 400 350 IBAT (mA) IBAT (mA) 300 250 200 150 100 50 0 2 2.5 3.5 3 VBAT (V) 4 4.5
3558 G05
Battery Charge Current vs Ambient Temperature in Thermal Regulation
500 450 400 350 300 250 200 150 VCC = 5V HPWR = 5V RPROG = 1.74k EN1 = EN2 = 0 5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G06
VCC = 5V RPROG = 1.74k
HPWR = 5V
475
HPWR = 0V
100 50
0 -55 -35 -15
Battery Charger Undervoltage Lockout Threshold vs Temperature
4.2 4.1 4.0 VCC (V) 3.9 3.8 3.7 3.6 3.5 -55 -35 IBAT (A) BAT = 3.5V RISING 3.0 2.5
Battery Drain Current in Undervoltage Lockout vs Temperature
1.2 EN1 = EN2 = 0V 1.0 BAT = 4.2V 2.0 0.8 BAT = 3.6V 1.5 1.0 0.5 0 -55 -35 VPROG (V) 0.6 0.4 0.2 0 25 5 45 -15 TEMPERATURE (C) 65 85
3558 G08
PROG Voltage vs Battery Charge Current
VCC = 5V HPWR = 5V RPROG = 1.74k EN1 = EN2 = 0V
FALLING
25 5 45 -15 TEMPERATURE (C)
65
85
3558 G07
0
50 100 150 200 250 300 350 400 450 500 IBAT (mA)
3558 G09
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LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS
Recharge Threshold vs Temperature
115 111 107 VRECHARGE (mV) 103 RDS(ON) (m) 99 95 91 87 83 79 75 -55 -35 25 5 -15 45 TEMPERATURE (C) 65 85 400 350 300 -55 -35 -15 5 25 45 65 85 0.6 0.5 0.4 -55 -35 -15 45 25 5 TEMPERATURE (C) 65 85
3558 G12
TA = 25C, unless otherwise noted. SUSP/HPWR Pin Rising Thresholds vs Temperature
1.2 VCC = 5V 1.1 1.0 THRESHOLD (V) 0.9 0.8 0.7
Battery Charger FET On-Resistance vs Temperature
700 650 600 550 500 450 VCC = 4V IBAT = 200mA EN1 = EN2 = 0V
VCC = 5V
TEMPERATURE (C)
3558 G10 3558 G11
CHRG Pin Output Low Voltage vs Temperature
140 120 100 VOLTAGE (mV) 80 60 40 20 0 -55 -35 ICHRG (mA) VCC = 5V ICHRG = 5mA 70 60 50 40 30 20 10 25 5 45 -15 TEMPERATURE (C) 65 85
3558 G13
CHRG Pin I-V Curve
2.0 VCC = 5V BAT = 3.8V PERCENT ERROR (%) 1.5 1.0 0.5 0 -0.5 -1.0 0 1 2 4 3 CHRG (V) 5 6
3558 G14
Timer Accuracy vs Supply Voltage
0
4.3
4.5
4.7
4.9 VCC (V)
5.1
5.3
5.5
3558 G15
Timer Accuracy vs Temperature
7 6 5 PERCENT ERROR (%) 4 3 2 1 0 -1 -2 -55 -35 -15 5 25 45 TEMPERATURE (C) 65 85 CHRG (V) BAT (V) IBAT (mA) VCC = 5V 1000 800 600 400 200 0 5.0 4.5 4.0 3.5 3.0 5.0 4.0 3.0 2.0 1.0 0
Complete Charge Cycle 2400mAh Battery
2.425 VCC = 5V RPROG = 0.845k HPWR = 5V FREQUENCY (MHz) 2.325 2.225
Buck and Buck-Boost Regulator Switching Frequency vs Temperature
VCC = 0V, MODE = 0 BAT = PVIN1 = PVIN2 BAT = 4.2V BAT = 2.7V 2.125 2.025 1.925 1.825 1.725 -55 -35 -15 BAT = 3.6V
0
1
2
4 3 TIME (HOUR)
5
6
3558 G17
5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G18
3558 G16
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LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS
Buck and Buck-Boost Regulator Undervoltage Thresholds vs Temperature
2.750 2.700 2.650 INPUT VOLTAGE (V) 2.600 2.550 2.500 2.450 2.400 2.350 2.300 2.250 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G19
TA = 25C, unless otherwise noted.
Buck and Buck-Boost Regulator Enable Thresholds vs Temperature
1200 1100 INPUT CURRENT (A) 1000 BAT = PVIN1 = PVIN2 = 3.6V 50 45 40
Buck Regulator Input Current vs Temperature, Burst Mode Operation
FB1 = 0.85V
BAT = PVIN1 = PVIN2
RISING VEN (V)
900 800 700 600 500 400 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G20
PVIN1 = 4.2V 35 PVIN1 = 2.7V 30 25 20 -55 -35 -15
FALLING
RISING FALLING
5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G21
Buck Regulator Input Current vs Temperature, Pulse Skip Mode
400 350 1100 INPUT CURRENT (A) RDS(ON) (m) 300 PVIN1 = 4.2V 250 PVIN1 = 2.7V 200 150 100 -55 -35 -15 1000 900 800 700 600 500 5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G22
Buck Regulator PMOS RDS(0N) vs Temperature
1300 1300 1200 1100 RDS(ON) (m) 1000 900 800 700 600 500 5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G23
Buck Regulator NMOS RDS(0N) vs Temperature
FB1 = 0.85V 1200
PVIN1 = 2.7V
PVIN1 = 2.7V PVIN1 = 4.2V
PVIN1 = 4.2V
400 -55 -35 -15
400 -55 -35 -15
5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G24
Buck Regulator Efficiency vs ILOAD
100 90 80 EFFICIENCY (%) 70 PULSE SKIP MODE VOUT (V) 60 50 40 30 20 10 0 0.1 1 10 ILOAD (mA) VOUT = 1.2V PVIN1 = 2.7V PVIN1 = 4.2V 100 1000
3558 G25
Buck Regulator Load Regulation
PVIN1 = 3.6V 1.24 VOUT = 1.2V 1.23 1.22 1.21 1.20 1.19 1.18 1.17 1.16 1.15 1 10 ILOAD (mA)
3558 G26
Buck Regulator Line Regulation
1.250 1.240 1.230 ILOAD = 200mA
1.25
Burst Mode OPERATION
Burst Mode OPERATION VOUT (V)
1.220 1.210 1.200 1.190 1.180 1.170 1.160 100 1000 1.150 2.700 3.000 3.600 3.300 PVIN1 (V) 3.900 4.200
3558 G27
PULSE SKIP MODE
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LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS
Buck Regulator Start-Up Transient
VOUT 500mV/DIV VOUT 20mV/ DIV (AC) SW 2V/DIV INDUCTOR CURRENT IL = 50mA/ DIV PVIN1 = 3.8V 50s/DIV PULSE SKIP MODE LOAD = 6
3558 G28
TA = 25C, unless otherwise noted. Buck Regulator Burst Mode Operation
VOUT 20mV/ DIV (AC) SW 2V/DIV INDUCTOR CURRENT IL = 60mA/ DIV
Buck Regulator Pulse Skip Mode Operation
INDUCTOR CURRENT IL = 200mA/ DIV EN 2V/DIV
PVIN1 = 3.8V LOAD = 10mA
200ns/DIV
3558 G29
PVIN1 = 3.8V LOAD = 60mA
2s/DIV
3558 G30
Buck Regulator Transient Response, Pulse Skip Mode
INDUCTOR CURRENT IL = 200mA/ DIV INDUCTOR CURRENT IL = 200mA/ DIV
Buck Regulator Transient Response, Burst Mode Operation
30
Buck-Boost Regulator Input Current vs Temperature
Burst Mode OPERATION FB2 = 0.85V
25 INPUT CURRENT (A) PVIN2 = 4.2V 20 PVIN2 = 2.7V 15
VOUT 50mV/ DIV (AC) LOAD STEP 5mA TO 290mA PVIN1 = 3.8V 50s/DIV
3558 G31
VOUT 50mV/ DIV (AC) LOAD STEP 5mA TO 290mA PVIN1 = 3.8V 50s/DIV
3558 G32
10
5 -55 -35 -15
5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G33
Buck-Boost Regulator Input Current vs Temperature
500 PWM MODE 450 FB2 = 0.85V INPUT CURRENT (A) 400 RDS(ON) (m) 350 300 250 200 150 100 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G34
Buck-Boost Regulator PMOS RDS(ON) vs Temperature
800 750 700 650 600 550 500 450 400 350 300 250 200 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G35
Buck-Boost Regulator NMOS RDS(ON) vs Temperature
1200 1100 1000
PVIN2 = 2.7V RDS(ON) (m)
900 800 700 600 500 400 300 200 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
3558 G36
PVIN2 = 2.7V
PVIN2 = 4.2V PVIN2 = 2.7V
PVIN2 = 4.2V
PVIN2 = 4.2V
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LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS
Buck-Boost Regulator Efficiency vs Input Voltage
100 95 90 EFFICIENCY (%) EFFICIENCY (%) 85 80 75 70 65 60 55 50 2.700 3.000 3.600 3.300 PVIN2 (V) Burst Mode OPERATION PWM MODE 3.900 4.200
3558 G37
TA = 25C, unless otherwise noted.
Buck-Boost Efficiency vs Load Current
100 VOUT = 3.3V 3.6V 4.2V 2.7V 90 80 70 60 50 40 30 20 10 0 0.10 2.7V 3.6V 4.2V 1 PVIN2, Burst Mode OPERATION PVIN2, PWM MODE 10 ILOAD (mA) 100 1000
3558 G38
VOUT = 3.3V
ILOAD = 10mA
ILOAD = 1mA
ILOAD = 100mA ILOAD = 400mA
Buck-Boost Regulator Load Regulation
3.36 3.35 3.34 3.33 3.32 VOUT (V) 3.31 3.30 3.29 3.28 3.27 3.26 3.25 3.24 0.10 1 10 ILOAD (mA) 100 1000
3558 G39
Buck-Boost Regulator Line Regulation
3.36 3.35 3.34 3.33
PVIN2 = 3.6V
Burst Mode OPERATION PWM MODE VOUT (V)
3.32 3.31 3.30 3.29 3.28 3.27 3.26 3.25 3.24 2.700 3.000 3.600 3.300 PVIN2 (V) 3.900 4.200
3558 G40
PWM MODE ILOAD = 100mA Burst Mode OPERATION ILOAD = 10mA
Buck-Boost Regulator Start-Up Transient, Burst Mode Operation
PVIN2 = 3.6V RLOAD = 332
Buck-Boost Regulator Start-Up Transient, PWM Mode
PVIN2 = 3.6V RLOAD = 16
VOUT 1V/DIV INDUCTOR CURRENT IL = 200mA/DIV EN2 1V/DIV 100s/DIV
3558 G41
VOUT 1V/DIV INDUCTOR CURRENT IL = 200mA/DIV EN2 1V/DIV 100s/DIV
3558 G42
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LTC3558 PIN FUNCTIONS
GND (Pin 1): Ground. Connect to Exposed Pad (Pin 21). BAT (Pin 2): Charge Current Output. Provides charge current to the battery and regulates final float voltage to 4.2V. MODE (Pin 3): MODE Pin for Switching Regulators. When held high, both regulators operate in Burst Mode Operation. When held low, the buck regulator operates in pulse skip mode and the buck-boost regulator operates in PWM mode. This pin is a high impedance input; do not float. FB1 (Pin 4): Buck Regulator Feedback Voltage Pin. Receives feedback by a resistor divider connected across the output. EN1 (Pin 5): Enable Input Pin for the Buck Regulator. This pin is a high impedance input; do not float. Active high. SW1 (Pin 6): Buck Regulator Switching Node. External inductor connects to this node. PVIN1 (Pin 7): Input Supply Pin for Buck Regulator. Connect to BAT and PVIN2. A single 10F input decoupling capacitor to GND is required. PVIN2 (Pin 8): Input Supply Pin for Buck-Boost Regulator. Connect to BAT and PVIN1. A single 10F input decoupling capacitor to GND is required. SWAB2 (Pin 9): Switch Node for Buck-Boost Regulator Connected to the Internal Power Switches A and B. External inductor connects between this node and SWCD2. SWCD2 (Pin 10): Switch Node for Buck-Boost Regulator Connected to the Internal Power Switches C and D. External inductor connects between this node and SWAB2. VOUT2 (Pin 11): Regulated Output Voltage for Buck-Boost Regulator. SUSP (Pin 12): Suspend Battery Charging Operation. A voltage greater than 1.2V on this pin puts the battery charger in suspend mode, disables the charger and resets the termination timer. A weak pull-down current is internally applied to this pin to ensure it is low at power-up when the input is not being driven externally. FB2 (Pin 13): Buck-Boost Regulator Feedback Voltage Pin. Receives feedback by a resistor divider connected across the output. VC2 (Pin 14): Output of the Error Amplifier and Voltage Compensation Node for the Buck-Boost Regulator. External Type I or Type III compensation (to FB2) connects to this pin. EN2 (Pin 15): Enable Input Pin for the Buck-Boost Regulator. This pin is a high impedance input; do not float. Active high. HPWR (Pin 16): High Current Battery Charging Enabled. A voltage greater than 1.2V at this pin programs the BAT pin current at 100% of the maximum programmed charge current. A voltage less than 0.4V sets the BAT pin current to 20% of the maximum programmed charge current. When used with a 1.74k PROG resistor, this pin can toggle between low power and high power modes per USB specification. A weak pull-down current is internally applied to this pin to ensure it is low at power-up when the input is not being driven externally. NTC (Pin 17): Input to the NTC Thermistor Monitoring Circuit. The NTC pin connects to a negative temperature coefficient thermistor which is typically co-packaged with the battery pack to determine if the battery is too hot or too cold to charge. If the battery temperature is out of range, charging is paused until the battery temperature re-enters the valid range. A low drift bias resistor is required from VCC to NTC and a thermistor is required from NTC to ground. To disable the NTC function, the NTC pin should be tied to ground. PROG (Pin 18): Charge Current Program and Charge Current Monitor Pin. Charge current is programmed by connecting a resistor from PROG to ground. When charging in constant-current mode, the PROG pin servos to 1V if the HPWR pin is pulled high, or 200mV if the HPWR pin is pulled low. The voltage on this pin always represents the BAT pin current through the following formula: IBAT = PROG * 800 RPROG
CHRG (Pin 19): Open-Drain Charge Status Output. The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG charging, not charging (i.e., the charge current is less than one-tenth
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LTC3558 PIN FUNCTIONS
of the full-scale charge current), unresponsive battery (i.e., the battery voltage remains below 2.9V after one half hour of charging) and battery temperature out of range. CHRG requires a pull-up resistor and/or LED to provide indication. VCC (Pin 20): Battery Charger Input. A 1F decoupling capacitor to GND is recommended. Exposed Pad (Pin 21): Ground. The Exposed Pad must be soldered to PCB ground to provide electrical contact and rated thermal performance.
BLOCK DIAGRAM
20 VCC
VCC
BAT BODY MAXER
1x
800x BAT
2
19 16 12
CHRG HPWR SUSP LOGIC CA TA TDIE PROG NTCA 18
17
NTC NTC REF MODE EN1 EN2 FB1 OT TDIE 0.8V UNDERVOLTAGE LOCKOUT CLK CONTROL LOGIC Gm MN EN MODE MP SW1 6 PVIN1 PVIN2 PVIN1 7 BATTERY CHARGER
3 5 15 4
BANDGAP
VREF = 0.8V
OSCILLATOR 2.25MHz 13 FB2
CLK
BUCK-BOOST REGULATOR EN CLK MODE AD SWAB2 CONTROL LOGIC BC SWCD2 9 10
ERROR AMP
0.8V 14 VC2
GND 1
12
+
DIE TEMPERATURE
BUCK REGULATOR PVIN2 VOUT2 8 11
VC2
EXPOSED PAD 21
3558 BD
+
3558f
- - + -
LTC3558 OPERATION
The LTC3558 is a linear battery charger with a monolithic synchronous buck regulator and a monolithic synchronous buck-boost regulator. The buck regulator is internally compensated and needs no external compensation components. The battery charger employs a constant-current, constantvoltage charging algorithm and is capable of charging a single Li-Ion battery at charging currents up to 950mA. The user can program the maximum charging current available at the BAT pin via a single PROG resistor. The actual BAT pin current is set by the status of the HPWR pin. For proper operation, the BAT, PVIN1 and PVIN2 pins must be tied together, as shown in Figure 1. Current being delivered at the BAT pin is 500mA. Both switching regulators are enabled. The sum of the average input currents drawn by both switching regulators is 200mA. This makes the effective battery charging current only 300mA. If the HPWR pin were tied LO, the BAT pin current would be 100mA. With the switching regulator conditions unchanged, this would cause the battery to discharge at 100mA.
500mA USB (5V) VCC PROG RPROG SUSP HIGH HIGH HIGH LOW HPWR EN1 EN2 MODE SWCD2 VOUT2 SW1
3558 F01
300mA 200mA
BAT PVIN1 PVIN2 LTC3558 SWAB2 2.2H
+
10F
SINGLE Li-lon CELL 3.6V
+
VOUT1
Figure 1. For Proper Operation, the BAT, PVIN1 and PVIN2 Pins Must Be Tied Together
APPLICATIONS INFORMATION
Battery Charger Introduction The LTC3558 has a linear battery charger designed to charge single-cell lithium-ion batteries. The charger uses a constant-current/constant-voltage charge algorithm with a charge current programmable up to 950mA. Additional features include automatic recharge, an internal termination timer, low-battery trickle charge conditioning, bad-battery detection, and a thermistor sensor input for out of temperature charge pausing. Furthermore, the battery charger is capable of operating from a USB power source. In this application, charge current can be programmed to a maximum of 100mA or 500mA per USB power specifications. Input Current vs Charge Current The battery charger regulates the total current delivered to the BAT pin; this is the charge current. To calculate the total input current (i.e., the total current drawn from the VCC pin), it is necessary to sum the battery charge current, charger quiescent current and PROG pin current. Undervoltage Lockout (UVLO) The undervoltage lockout circuit monitors the input voltage (VCC) and disables the battery charger until VCC rises above VUVLO (typically 4V). 200mV of hysteresis prevents oscillations around the trip point. In addition, a differential undervoltage lockout circuit disables the battery charger
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LTC3558 APPLICATIONS INFORMATION
when VCC falls to within VDUVLO (typically 50mV) of the BAT voltage. Suspend Mode The battery charger can also be disabled by pulling the SUSP pin above 1.2V. In suspend mode, the battery drain current is reduced to 1.5A and the input current is reduced to 8.5A. Charge Cycle Overview When a battery charge cycle begins, the battery charger first determines if the battery is deeply discharged. If the battery voltage is below VTRKL, typically 2.9V, an automatic trickle charge feature sets the battery charge current to 10% of the full-scale value. Once the battery voltage is above 2.9V, the battery charger begins charging in constant-current mode. When the battery voltage approaches the 4.2V required to maintain a full charge, otherwise known as the float voltage, the charge current begins to decrease as the battery charger switches into constant-voltage mode. Trickle Charge and Defective Battery Detection Any time the battery voltage is below VTRKL, the charger goes into trickle charge mode and reduces the charge current to 10% of the full-scale current. If the battery voltage remains below VTRKL for more than 1/2 hour, the charger latches the bad-battery state, automatically terminates, and indicates via the CHRG pin that the battery was unresponsive. If for any reason the battery voltage rises above VTRKL, the charger will resume charging. Since the charger has latched the bad-battery state, if the battery voltage then falls below VTRKL again but without rising past VRECHRG first, the charger will immediately assume that the battery is defective. To reset the charger (i.e., when the dead battery is replaced with a new battery), simply remove the input voltage and reapply it or put the part in and out of suspend mode. Charge Termination The battery charger has a built-in safety timer that sets the total charge time for 4 hours. Once the battery voltage rises above VRECHRG (typically 4.105V) and the charger enters constant-voltage mode, the 4-hour timer is started. After the safety timer expires, charging of the battery will discontinue and no more current will be delivered. Automatic Recharge After the battery charger terminates, it will remain off, drawing only microamperes of current from the battery. If the portable product remains in this state long enough, the battery will eventually self discharge. To ensure that the battery is always topped off, a charge cycle will automatically begin when the battery voltage falls below VRECHRG (typically 4.105V). In the event that the safety timer is running when the battery voltage falls below VRECHRG, it will reset back to zero. To prevent brief excursions below VRECHRG from resetting the safety timer, the battery voltage must be below VRECHRG for more than 1.7ms. The charge cycle and safety timer will also restart if the VCC UVLO or DUVLO cycles low and then high (e.g., VCC is removed and then replaced) or the charger enters and then exits suspend mode. Programming Charge Current The PROG pin serves both as a charge current program pin, and as a charge current monitor pin. By design, the PROG pin current is 1/800th of the battery charge current. Therefore, connecting a resistor from PROG to ground programs the charge current while measuring the PROG pin voltage allows the user to calculate the charge current. Full-scale charge current is defined as 100% of the constant-current mode charge current programmed by the PROG resistor. In constant-current mode, the PROG pin servos to 1V if HPWR is high, which corresponds to charging at the full-scale charge current, or 200mV if HPWR is low, which corresponds to charging at 20% of the fullscale charge current. Thus, the full-scale charge current and desired program resistor for a given full-scale charge current are calculated using the following equations: ICHG = 800 V RPROG 800 V ICHG
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RPROG =
14
LTC3558 APPLICATIONS INFORMATION
In any mode, the actual battery current can be determined by monitoring the PROG pin voltage and using the following equation: IBAT = PROG * 800 RPROG charge current has dropped to below 10% of the full-scale current, the CHRG pin is released (high impedance). If a fault occurs after the CHRG pin is released, the pin remains high impedance. However, if a fault occurs before the CHRG pin is released, the pin is switched at 35kHz. While switching, its duty cycle is modulated between a high and low value at a very low frequency. The low and high duty cycles are disparate enough to make an LED appear to be on or off thus giving the appearance of "blinking". Each of the two faults has its own unique "blink" rate for human recognition as well as two unique duty cycles for microprocessor recognition. Table 1 illustrates the four possible states of the CHRG pin when the battery charger is active.
Table 1. CHRG Output Pin
MODULATION (BLINK) FREQUENCY 0 Hz (Lo-Z) 0 Hz (Hi-Z) 1.5Hz at 50% 6.1Hz at 50%
Thermal Regulation To prevent thermal damage to the IC or surrounding components, an internal thermal feedback loop will automatically decrease the programmed charge current if the die temperature rises to approximately 115C. Thermal regulation protects the battery charger from excessive temperature due to high power operation or high ambient thermal conditions and allows the user to push the limits of the power handling capability with a given circuit board design without risk of damaging the LTC3558 or external components. The benefit of the LTC3558 battery charger thermal regulation loop is that charge current can be set according to actual conditions rather than worst-case conditions with the assurance that the battery charger will automatically reduce the current in worst-case conditions. Charge Status Indication The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG charging, not charging, unresponsive battery and battery temperature out of range. The signal at the CHRG pin can be easily recognized as one of the above four states by either a human or a microprocessor. The CHRG pin, which is an open-drain output, can drive an indicator LED through a current limiting resistor for human interfacing, or simply a pull-up resistor for microprocessor interfacing. To make the CHRG pin easily recognized by both humans and microprocessors, the pin is either a low for charging, a high for not charging, or it is switched at high frequency (35kHz) to indicate the two possible faults: unresponsive battery and battery temperature out of range. When charging begins, CHRG is pulled low and remains low for the duration of a normal charge cycle. When the
STATUS Charging IBAT < C/10 NTC Fault Bad Battery
FREQUENCY 0Hz 0Hz 35kHz 35kHz
DUTY CYCLE 100% 0% 6.25%, 93.75% 12.5%, 87.5%
An NTC fault is represented by a 35kHz pulse train whose duty cycle alternates between 6.25% and 93.75% at a 1.5Hz rate. A human will easily recognize the 1.5Hz rate as a "slow" blinking which indicates the out of range battery temperature while a microprocessor will be able to decode either the 6.25% or 93.75% duty cycles as an NTC fault. If a battery is found to be unresponsive to charging (i.e., its voltage remains below VTRKL for over 1/2 hour), the CHRG pin gives the battery fault indication. For this fault, a human would easily recognize the frantic 6.1Hz "fast" blinking of the LED while a microprocessor would be able to decode either the 12.5% or 87.5% duty cycles as a bad battery fault. Although very improbable, it is possible that a duty cycle reading could be taken at the bright-dim transition (low duty cycle to high duty cycle). When this happens the duty cycle reading will be precisely 50%. If the duty cycle reading is 50%, system software should disqualify it and take a new duty cycle reading.
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LTC3558 APPLICATIONS INFORMATION
NTC Thermistor The battery temperature is measured by placing a negative temperature coefficient (NTC) thermistor close to the battery pack. The NTC circuitry is shown in Figure 3. To use this feature, connect the NTC thermistor, RNTC, between the NTC pin and ground, and a bias resistor, RNOM, from VCC to NTC. RNOM should be a 1% resistor with a value equal to the value of the chosen NTC thermistor at 25C (R25). A 100k thermistor is recommended since thermistor current is not measured by the battery charger and its current will have to be considered for compliance with USB specifications. The battery charger will pause charging when the resistance of the NTC thermistor drops to 0.54 times the value of R25 or approximately 54k (for a Vishay "Curve 1" thermistor, this corresponds to approximately 40C). If the battery charger is in constant-voltage mode, the safety timer will pause until the thermistor indicates a return to a valid temperature. As the temperature drops, the resistance of the NTC thermistor rises. The battery charger is also designed to pause charging when the value of the NTC thermistor increases to 3.25 times the value of R25. For a Vishay "Curve 1" thermistor, this resistance, 325k, corresponds to approximately 0C. The hot and cold comparators each have approximately 3C of hysteresis to prevent oscillation about the trip point. Grounding the NTC pin disables all NTC functionality.
DUVLO, UVLO AND SUSPEND POWER ON NO IF SUSP < 0.4V AND VCC > 4V AND VCC > BAT + 130mV? YES FAULT NTC FAULT BATTERY CHARGING SUSPENDED CHRG PULSES NO FAULT BAT 2.9V BAT > 2.9V 2.9V < BAT < 4.105V
DISABLE MODE CHRG HIGH IMPEDANCE
STANDBY MODE NO CHARGE CURRENT CHRG HIGH IMPEDANCE
TRICKLE CHARGE MODE 1/10 FULL CHARGE CURRENT CHRG STRONG PULL-DOWN 30 MINUTE TIMER BEGINS 30 MINUTE TIMEOUT
CONSTANT CURRENT MODE FULL CHARGE CURRENT CHRG STRONG PULL-DOWN
4-HOUR TIMEOUT
DEFECTIVE BATTERY NO CHARGE CURRENT CHRG PULSES
CONSTANT VOLTAGE MODE 4-HOUR TERMINATION TIMER BEGINS
BAT DROPS BELOW 4.105V 4-HOUR TERMINATION TIMER RESETS
3558 F02
Figure 2. State Diagram of Battery Charger Operation
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LTC3558 APPLICATIONS INFORMATION
Alternate NTC Thermistors and Biasing The battery charger provides temperature qualified charging if a grounded thermistor and a bias resistor are connected to the NTC pin. By using a bias resistor whose value is equal to the room temperature resistance of the thermistor (R25) the upper and lower temperatures are pre-programmed to approximately 40C and 0C, respectively (assuming a Vishay "Curve 1" thermistor). The upper and lower temperature thresholds can be adjusted by either a modification of the bias resistor value or by adding a second adjustment resistor to the circuit. If only the bias resistor is adjusted, then either the upper or the lower threshold can be modified but not both. The other trip point will be determined by the characteristics of the thermistor. Using the bias resistor in addition to an adjustment resistor, both the upper and the lower temperature trip points can be independently programmed with the constraint that the difference between the upper and lower temperature thresholds cannot decrease. Examples of each technique are given below. NTC thermistors have temperature characteristics which are indicated on resistance-temperature conversion tables. The Vishay-Dale thermistor NTHS0603N011-N1003F used , in the following examples, has a nominal value of 100k and follows the Vishay "Curve 1" resistance-temperature characteristic. In the explanation below, the following notation is used. R25 = Value of the thermistor at 25C RNTC|COLD = Value of thermistor at the cold trip point RNTC|HOT = Value of the thermistor at the hot trip point rCOLD = Ratio of RNTC|COLD to R25 rHOT = Ratio of RNTC|HOT to R25 RNOM = Primary thermistor bias resistor (see Figure 3) R1 = Optional temperature range adjustment resistor (see Figure 4) The trip points for the battery charger's temperature qualification are internally programmed at 0.349 * VCC for the hot threshold and 0.765 * VCC for the cold threshold. Therefore, the hot trip point is set when: RNTCHOT | RNOM + RNTCHOT | * VCC = 0.349 * VCC
and the cold trip point is set when: RNTC|COLD RNOM + RNTC|COLD * VCC = 0.765 * VCC
Solving these equations for RNTC|COLD and RNTC|HOT results in the following: RNTC|HOT = 0.536 * RNOM and RNTC|COLD = 3.25 * RNOM By setting RNOM equal to R25, the above equations result in rHOT = 0.536 and rCOLD = 3.25. Referencing these ratios to the Vishay Resistance-Temperature Curve 1 chart gives a hot trip point of about 40C and a cold trip point of about 0C. The difference between the hot and cold trip points is approximately 40C. By using a bias resistor, RNOM, different in value from R25, the hot and cold trip points can be moved in either direction. The temperature span will change somewhat due to the nonlinear behavior of the thermistor. The following equations can be used to easily calculate a new value for the bias resistor: r RNOM = HOT * R25 0.536 RNOM = rCOLD * R25 3.25
where rHOT and rCOLD are the resistance ratios at the desired hot and cold trip points. Note that these equations are linked. Therefore, only one of the two trip points can be chosen, the other is determined by the default ratios designed in the IC. Consider an example where a 60C hot trip point is desired. From the Vishay Curve 1 R-T characteristics, rHOT is 0.2488 at 60C. Using the above equation, RNOM should be set
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LTC3558 APPLICATIONS INFORMATION
to 46.4k. With this value of RNOM, the cold trip point is about 16C. Notice that the span is now 44C rather than the previous 40C. The upper and lower temperature trip points can be independently programmed by using an additional bias resistor as shown in Figure 4. The following formulas can be used to compute the values of RNOM and R1: RNOM = rCOLD - rHOT * R25 2.714 For example, to set the trip points to 0C and 45C with a Vishay Curve 1 thermistor choose: RNOM = 3.266 - 0.4368 * 100k = 104.2k 2.714
the nearest 1% value is 105k. R1 = 0.536 * 105k - 0.4368 * 100k = 12.6k the nearest 1% value is 12.7k. The final solution is shown in Figure 4 and results in an upper trip point of 45C and a lower trip point of 0C.
R1 = 0.536 * RNOM - rHOT * R25
20 RNOM 100k 17 RNTC 100k
VCC
NTC BLOCK 0.765 * VCC (NTC RISING)
20 RNOM 105k TOO_COLD 17 R1 12.7k RNTC 100k
VCC 0.765 * VCC (NTC RISING)
+
NTC_ENABLE 0.017 * VCC (NTC FALLING)
-
3558 F03
0.017 * VCC (NTC FALLING)
-
3558 F04
Figure 3. Typical NTC Thermistor Circuit
Figure 4. NTC Thermistor Circuit with Additional Bias Resistor
18
+
+
0.349 * VCC (NTC FALLING)
TOO_HOT
0.349 * VCC (NTC FALLING)
+ -
TOO_HOT
+ -
NTC
NTC
-
TOO_COLD
-
+
NTC_ENABLE
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LTC3558 APPLICATIONS INFORMATION
USB and Wall Adapter Power Although the battery charger is designed to draw power from a USB port to charge Li-Ion batteries, a wall adapter can also be used. Figure 5 shows an example of how to combine wall adapter and USB power inputs. A P-channel MOSFET, MP1, is used to prevent back conduction into the USB port when a wall adapter is present and Schottky diode, D1, is used to prevent USB power loss through the 1k pull-down resistor. Typically, a wall adapter can supply significantly more current than the 500mA-limited USB port. Therefore, an N-channel MOSFET, MN1, and an extra program resistor are used to increase the maximum charge current to 950mA when the wall adapter is present. current. It is not necessary to perform any worst-case power dissipation scenarios because the LTC3558 will automatically reduce the charge current to maintain the die temperature at approximately 105C. However, the approximate ambient temperature at which the thermal feedback begins to protect the IC is: TA = 105C - PD JA TA = 105C - ( VCC - VBAT ) * IBAT * JA Example: Consider an LTC3558 operating from a USB port providing 500mA to a 3.5V Li-Ion battery. The ambient temperature above which the LTC3558 will begin to reduce the 500mA charge current is approximately: TA = 105C - ( 5V - 3.5V ) * ( 500mA ) * 68C / W TA = 105C - 0.75W * 68C / W = 105C - 51C TA = 54C The LTC3558 can be used above 70C, but the charge current will be reduced from 500mA. The approximate current at a given ambient temperature can be calculated: IBAT =
3558 F05
5V WALL ADAPTER 950mA ICHG USB POWER 500mA ICHG
IBAT D1 BAT BATTERY CHARGER VCC PROG MN1 1.65k 1k
MP1
+
Li-Ion BATTERY
1.74k
105C - TA ( VCC - VBAT ) * JA
Figure 5. Combining Wall Adapter and USB Power
Using the previous example with an ambient temperature of 88C, the charge current will be reduced to approximately: IBAT = 105C - 88C 17C = (5V - 3.5V ) * 68C / W 102C / A
Power Dissipation The conditions that cause the LTC3558 to reduce charge current through thermal feedback can be approximated by considering the power dissipated in the IC. For high charge currents, the LTC3558 power dissipation is approximately: PD = ( VCC - VBAT ) * IBAT where PD is the power dissipated, VCC is the input supply voltage, VBAT is the battery voltage, and IBAT is the charge
IBAT = 167mA Furthermore, the voltage at the PROG pin will change proportionally with the charge current as discussed in the Programming Charge Current section. It is important to remember that LTC3558 applications do not need to be designed for worst-case thermal conditions since the IC will automatically reduce power dissipation when the junction temperature reaches approximately 105C.
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LTC3558 APPLICATIONS INFORMATION
Battery Charger Stability Considerations The LTC3558 battery charger contains two control loops: the constant-voltage and constant-current loops. The constantvoltage loop is stable without any compensation when a battery is connected with low impedance leads. Excessive lead length, however, may add enough series inductance to require a bypass capacitor of at least 1.5F from BAT to GND. Furthermore, a 4.7F capacitor with a 0.2 to 1 series resistor from BAT to GND is required to keep ripple voltage low when the battery is disconnected. High value capacitors with very low ESR (especially ceramic) reduce the constant-voltage loop phase margin, possibly resulting in instability. Ceramic capacitors up to 22F may be used in parallel with a battery, but larger ceramics should be decoupled with 0.2 to 1 of series resistance. In constant-current mode, the PROG pin is in the feedback loop, not the battery. Because of the additional pole created by the PROG pin capacitance, capacitance on this pin must be kept to a minimum. With no additional capacitance on the PROG pin, the charger is stable with program resistor values as high as 25K. However, additional capacitance on this node reduces the maximum allowed program resistor. The pole frequency at the PROG pin should be kept above 100kHz. Therefore, if the PROG pin is loaded with a capacitance, CPROG, the following equation should be used to calculate the maximum resistance value for RPROG: RPROG 1 2 * 10 * CPROG
MP1 Si2333 VCC LTC3558 10k PROG GND RPROG
3558 F06 3558 F07
Average, rather than instantaneous, battery current may be of interest to the user. For example, if a switching power supply operating in low-current mode is connected in parallel with the battery, the average current being pulled out of the BAT pin is typically of more interest than the instantaneous current pulses. In such a case, a simple RC filter can be used on the PROG pin to measure the average battery current as shown in Figure 6. A 10k resistor has been added between the PROG pin and the filter capacitor to ensure stability. USB Inrush Limiting When a USB cable is plugged into a portable product, the inductance of the cable and the high-Q ceramic input capacitor form an L-C resonant circuit. If there is not much impedance in the cable, it is possible for the voltage at the input of the product to reach as high as twice the USB voltage (~10V) before it settles out. In fact, due to the high voltage coefficient of many ceramic capacitors (a nonlinearity), the voltage may even exceed twice the USB voltage. To prevent excessive voltage from damaging the LTC3558 during a hot insertion, the soft connect circuit in Figure 7 can be employed. In the circuit of Figure 7, capacitor C1 holds MP1 off when the cable is first connected. Eventually C1 begins to charge up to the USB input voltage applying increasing gate support to MP1. The long time constant of R1 and C1 prevents the current from building up in the cable too fast thus dampening out any resonant overshoot.
5
CFILTER
CHARGE CURRENT MONITOR CIRCUITRY
5V USB INPUT
C1 100nF USB CABLE R1 40k C2 10F LTC3558
GND
Figure 6. Isolated Capacitive Load on PROG Pin and Filtering
Figure 7. USB Soft Connect Circuit
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LTC3558 APPLICATIONS INFORMATION
Buck Switching Regulator General Information The LTC3558 contains a 2.25MHz constant-frequency current mode buck switching regulator that can provide up to 400mA. The switcher can be programmed for a minimum output voltage of 0.8V and can be used to power a microcontroller core, microcontroller I/O, memory or other logic circuitry. The regulator supports 100% duty cycle operation (dropout mode) when the input voltage drops very close to the output voltage and is also capable of operating in Burst Mode operation for highest efficiencies at light loads (Burst Mode operation is pin selectable). The buck switching regulator also includes soft-start to limit inrush current when powering on, short-circuit current protection, and switch node slew limiting circuitry to reduce radiated EMI. A MODE pin sets the buck switching regulator in Burst Mode operation or pulse skip operating mode. The regulator is enabled individually through its enable pin. The buck regulator input supply (PVIN1) should be connected to the battery pin (BAT) and PVIN2. This allows the undervoltage lockout circuit on the BAT pin to disable the buck regulators when the BAT voltage drops below 2.45V. Do not drive the buck switching regulator from a voltage other than BAT. A 10F decoupling capacitor from the PVIN1 pin to GND is recommended. Buck Switching Regulator Output Voltage Programming The buck switching regulator can be programmed for output voltages greater than 0.8V. The output voltage for the buck switching regulator is programmed using a resistor divider from the switching regulator output connected to its feedback pin (FB1), as shown in Figure 8, such that: VOUT = 0.8(1 + R1/R2) Typical values for R1 are in the range of 40k to 1M. The capacitor CFB cancels the pole created by feedback resistors and the input capacitance of the FB pin and also helps to improve transient response for output voltages much greater than 0.8V. A variety of capacitor sizes can be used for CFB but a value of 10pF is recommended for most applications. Experimentation with capacitor sizes between 2pF and 22pF may yield improved transient response if so desired by the user. Buck Switching Regulator Operating Modes The buck switching regulator includes two possible operating modes to meet the noise/power needs of a variety of applications. In pulse skip mode, an internal latch is set at the start of every cycle, which turns on the main P-channel MOSFET
PVIN EN PWM CONTROL MODE MP SW L VOUT CFB R1 FB 0.8V GND R2 CO
MN
3558 F08
Figure 8. Buck Converter Application Circuit
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LTC3558 APPLICATIONS INFORMATION
switch. During each cycle, a current comparator compares the peak inductor current to the output of an error amplifier. The output of the current comparator resets the internal latch, which causes the main P-channel MOSFET switch to turn off and the N-channel MOSFET synchronous rectifier to turn on. The N-channel MOSFET synchronous rectifier turns off at the end of the 2.25MHz cycle or if the current through the N-channel MOSFET synchronous rectifier drops to zero. Using this method of operation, the error amplifier adjusts the peak inductor current to deliver the required output power. All necessary compensation is internal to the buck switching regulator requiring only a single ceramic output capacitor for stability. At light loads in pulse skip mode, the inductor current may reach zero on each pulse which will turn off the N-channel MOSFET synchronous rectifier. In this case, the switch node (SW1) goes high impedance and the switch node voltage will "ring". This is discontinuous operation, and is normal behavior for a switching regulator. At very light loads in pulse skip mode, the buck switching regulator will automatically skip pulses as needed to maintain output regulation. At high duty cycle (VOUT > PVIN1 /2) in pulse skip mode, it is possible for the inductor current to reverse causing the buck converter to switch continuously. Regulation and low noise operation are maintained but the input supply current will increase to a couple mA due to the continuous gate switching. During Burst Mode operation, the buck switching regulator automatically switches between fixed frequency PWM operation and hysteretic control as a function of the load current. At light loads the buck switching regulator controls the inductor current directly and use a hysteretic control loop to minimize both noise and switching losses. During Burst Mode operation, the output capacitor is charged to a voltage slightly higher than the regulation point. The buck switching regulator then goes into sleep mode, during which the output capacitor provides the load current. In sleep mode, most of the switching regulator's circuitry is powered down, helping conserve battery power. When the output voltage drops below a pre-determined value, the buck switching regulator circuitry is powered on and another burst cycle begins. The sleep time decreases as the load current increases. Beyond a certain load current point (about 1/4 rated output load current) the buck switching regulator will switch to a low noise constant-frequency PWM mode of operation, much the same as pulse skip operation at high loads. For applications that can tolerate some output ripple at low output currents, Burst Mode operation provides better efficiency than pulse skip at light loads. The buck switching regulator allows mode transition onthe-fly, providing seamless transition between modes even under load. This allows the user to switch back and forth between modes to reduce output ripple or increase low current efficiency as needed. Burst Mode operation is set by driving the MODE pin high, while pulse skip mode is achieved by driving the MODE pin low. Buck Switching Regulator in Shutdown The buck switching regulator is in shutdown when not enabled for operation. In shutdown, all circuitry in the buck switching regulator is disconnected from the regulator input supply, leaving only a few nanoamps of leakage pulled to ground through a 13k resistor on the switch (SW1) pin when in shutdown. Buck Switching Regulator Dropout Operation It is possible for the buck switching regulator's input voltage to approach its programmed output voltage (e.g., a battery voltage of 3.4V with a programmed output voltage of 3.3V). When this happens, the PMOS switch duty cycle increases until it is turned on continuously at 100%. In this dropout condition, the respective output voltage equals the regulator's input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor.
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LTC3558 APPLICATIONS INFORMATION
Buck Switching Regulator Soft-Start Operation Soft-start is accomplished by gradually increasing the peak inductor current for each switching regulator over a 500s period. This allows an output to rise slowly, helping minimize the battery in-rush current required to charge up the regulator's output capacitor. A soft-start cycle occurs when the buck switcher first turns on, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes using the MODE pin. This allows seamless output operation when transitioning between operating modes. Buck Switching Regulator Switching Slew Rate Control The buck switching regulator contains circuitry to limit the slew rate of the switch node (SW1). This circuitry is designed to transition the switch node over a period of a couple of nanoseconds, significantly reducing radiated EMI and conducted supply noise while maintaining high efficiency. Buck Switching Regulator Low Supply Operation An undervoltage lockout (UVLO) circuit on PVIN1 shuts down the step-down switching regulators when BAT drops below 2.45V. This UVLO prevents the buck switching regulator from operating at low supply voltages where loss of regulation or other undesirable operation may occur. Buck Switching Regulator Inductor Selection The buck switching regulator is designed to work with inductors in the range of 2.2H to 10H, but for most applications a 4.7H inductor is suggested. Larger value inductors reduce ripple current which improves output ripple voltage. Lower value inductors result in higher ripple current which improves transient response time. To maximize efficiency, choose an inductor with a low DC resistance. For a 1.2V output efficiency is reduced about 2% for every 100m series resistance at 400mA load current, and about 2% for every 300m series resistance at 100mA load current. Choose an inductor with a DC current rating at least 1.5 times larger than the maximum load current to ensure that the inductor does not saturate during normal operation. If output short-circuit is a possible condition the inductor should be rated to handle the maximum peak current specified for the buck regulators. Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don't radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. Inductors that are very thin or have a very small volume typically have much higher DCR losses, and will not give the best efficiency. The choice of which style inductor to use often depends more on the price vs size, performance, and any radiated EMI requirements than on what the buck regulator requires to operate. The inductor value also has an effect on Burst Mode operation. Lower inductor values will cause Burst Mode switching frequency to increase.
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LTC3558 APPLICATIONS INFORMATION
Table 2 shows several inductors that work well with the LTC3558 buck switching regulator. These inductors offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information on their entire selection of inductors. Buck Switching Regulator Input/Output Capacitor Selection Low ESR (equivalent series resistance) ceramic capacitors should be used at switching regulator outputs as well as the switching regulator input supply. Ceramic capacitor dielectrics are a compromise between high dielectric constant and stability versus temperature and versus DC bias voltage. The X5R/X7R dielectrics offer the best compromise with high dielectric constant and acceptable performance over temperature and under bias. Do not use Y5V dielectrics. A 10F output capacitor is sufficient for most applications. For good transient response and stability the output capacitor should retain at least 4F of capacitance over operating temperature and bias voltage. The buck switching regulator input supply should be bypassed with a 10F capacitor. Consult manufacturer for detailed information on their selection and specifications of ceramic capacitors. Many manufacturers now offer very thin (< 1mm tall) ceramic capacitors ideal for use in height-restricted designs. Table 3 shows a list of several ceramic capacitor manufacturers.
Table 3: Recommended Ceramic Capacitor Manufacturers
AVX Murata Taiyo Yuden TDK (803) 448-9411 (714) 852-2001 (408) 537-4150 (888) 835-6646 www.avxcorp.com www.murata.com www.t-yuden.com www.tdk.com
Table 2. Recommended Inductors for Buck Switching Regulators
INDUCTOR TYPE DE2818C DE2812C CDRH3D16 SD3118 SD3112 LPS3015 *Typical DCR L (H) 4.7 4.7 4.7 4.7 4.7 4.7 MAX IDC (A) 1.25 1.15 0.9 1.3 0.8 1.1 MAX DCR (m) 72* 130* 110 162 246 200 SIZE IN mm (L x W x H) 3 x 2.8 x 1.8 3 x 2.8 x 1.2 4 x 4 x 1.8 3.1 x 3.1 x 1.8 3.1 x 3.1 x 1.2 3 x 3 x 1.5 MANUFACTURER Toko www.toko.com Sumida www.sumida.com Cooper www.cooperet.com Coilcraft www.coilcraft.com
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LTC3558 APPLICATIONS INFORMATION
Buck-Boost Switching Regulator The LTC3558 contains a 2.25MHz constant-frequency, voltage mode, buck-boost switching regulator. The regulator provides up to 400mA of output load current. The buck-boost switching regulator can be programmed for a minimum output voltage of 2.75V and can be used to power a microcontroller core, microcontroller I/O, memory, disk drive, or other logic circuitry. To suit a variety of applications, different mode functions allow the user to trade off noise for efficiency. Two modes are available to control the operation of the buck-boost regulator. At moderate to heavy loads, the constant-frequency PWM mode provides the least noise switching solution. At lighter loads, Burst Mode operation may be selected. Regulation is maintained by an error amplifier that compares the divided output voltage with a reference and adjusts the compensation voltage accordingly until the FB2 voltage has stabilized at 0.8V. The buck-boost switching regulator also includes soft-start to limit inrush current and voltage overshoot when powering on, short-circuit current protection, and switch node slew limiting circuitry for reduced radiated EMI. Buck-Boost Regulator PWM Operating Mode In PWM mode, the voltage seen at the feedback node is compared to a 0.8V reference. From the feedback voltage, an error amplifier generates an error signal seen at the VC2 pin. This error signal controls PWM waveforms that modulate switches A (input PMOS), B (input NMOS), C (output NMOS), and D (output PMOS). Switches A and B operate synchronously, as do switches C and D. If the input voltage is significantly greater than the programmed output voltage, then the regulator will operate in buck mode. In this case, switches A and B will be modulated, with switch D always on (and switch C always off), to stepdown the input voltage to the programmed output. If the input voltage is significantly less than the programmed output voltage, then the converter will operate in boost mode. In this case, switches C and D are modulated, with switch A always on (and switch B always off), to step up the input voltage to the programmed output. If the input voltage is close to the programmed output voltage, then the converter will operate in four-switch mode. While operating in four-switch mode, switches turn on as per the following sequence: switches A and D switches A and C switches B and D switches A and D. Buck-Boost Regulator Burst Mode Operation In Burst Mode operation, the switching regulator uses a hysteretic feedback voltage algorithm to control the output voltage. By limiting FET switching and using a hysteretic control loop switching losses are greatly reduced. In this mode, output current is limited to 50mA. While in Burst Mode operation, the output capacitor is charged to a voltage slightly higher than the regulation point. The buck-boost converter then goes into a SLEEP state, during which the output capacitor provides the load current. The output capacitor is charged by charging the inductor until the input current reaches 250mA typical, and then discharging the inductor until the reverse current reaches 0mA typical. This process of bursting current is repeated until the feedback voltage has charged to the reference voltage plus 6mV (806mV typical). In the SLEEP state, most of the regulator's circuitry is powered down, helping to conserve battery power. When the feedback voltage drops below the reference voltage minus 6mV (794mV typical), the switching regulator circuitry is powered on and another burst cycle begins. The duration for which the regulator operates in SLEEP depends on the load current and output capacitor value. The SLEEP time decreases as the load current increases. The maximum deliverable load current in Burst Mode operation is 50mA typical. The buck-boost regulator may not enter SLEEP if the load current is greater than 50mA. If the load current increases beyond this point while in Burst Mode operation, the output may lose regulation. Burst Mode operation provides a significant improvement in efficiency at light loads at the expense of higher output ripple when compared to PWM mode. For many noise-sensitive systems, Burst Mode operation might be undesirable at certain times (i.e., during a transmit or receive cycle of a wireless device), but highly desirable at others (i.e., when the device is in low power standby mode).
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LTC3558 APPLICATIONS INFORMATION
Buck-Boost Switching Regulator Output Voltage Programming The buck-boost switching regulator can be programmed for output voltages greater than 2.75V and less than 5.45V. To program the output voltage, a resistor divider is connected between VOUT2 and the feedback node (FB2) as shown in Figure 9. The output voltage is given by VOUT2 = 0.8(1 + R1/R2).
LTC3558 VOUT2 R1 FB2 R2
The output filter zero is given by: f FILTER _ ZERO = 1 2 * * RESR * COUT Hz
where RESR is the capacitor equivalent series resistance. A troublesome feature in boost mode is the right-half plane zero (RHP), and is given by: f RHPZ = PVIN22 Hz 2 * * IOUT * L * VOUT2
The loop gain is typically rolled off before the RHP zero frequency. A simple Type I compensation network, as shown in Figure 10, can be incorporated to stabilize the loop, but at the cost of reduced bandwidth and slower transient response. To ensure proper phase margin, the loop requires to be crossed over a decade before the LC double pole. The unity-gain frequency of the error amplifier with the Type I compensation is given by: f UG = 1 Hz 2 * * R1* CP1
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Figure 9. Programming the Buck-Boost Output Voltage Requires a Resistor Divider Connected Between VOUT2 and FB2
Closing the Feedback Loop The LTC3558 incorporates voltage mode PWM control. The control to output gain varies with operation region (buck, boost, buck-boost), but is usually no greater than 20. The output filter exhibits a double pole response given by: f FILTER _ POLE = 1 Hz 2 * * L * COUT
where COUT is the output filter capacitor.
VOUT2
+
ERROR AMP
0.8V FB2
R1
-
VC2 CP1 R2
3558 F10
Figure 10. Error Amplifier with Type I Compensation
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LTC3558 APPLICATIONS INFORMATION
Most applications demand an improved transient response to allow a smaller output filter capacitor. To achieve a higher bandwidth, Type III compensation is required. Two zeros are required to compensate for the double-pole response. Type III compensation also reduces any VOUT2 overshoot seen during a start-up condition. A Type III compensation circuit is shown in Figure 11 and yields the following transfer function: VC2 1 = VOUT 2 R1 (C1 + C2) * (1 + sR2C2) [1 + s (R1 + R3)C3 ] s 1 + sR2(C1|| C2) (1 + sR3C3) at the filter double pole. If they are placed at too low of a frequency, they will introduce too much gain to the system and the crossover frequency will be too high. The two high frequency poles should be placed such that the system crosses unity gain during the phase bump introduced by the zeros and before the boost right-half plane zero and such that the compensator bandwidth is less than the bandwidth of the error amp (typically 900kHz). If the gain of the compensation network is ever greater than the gain of the error amplifier, then the error amplifier no longer acts as an ideal op amp, and another pole will be introduced at the same point. Recommended Type III compensation components for a 3.3V output are: R1: 324k RFB: 105k C1: 10pF R2: 15k C2: 330pF R3: 121k C3: 33pF COUT : 22F LOUT : 2.2H
A Type III compensation network attempts to introduce a phase bump at a higher frequency than the LC double pole. This allows the system to cross unity gain after the LC double pole, and achieve a higher bandwidth. While attempting to cross over after the LC double pole, the system must still cross over before the boost right-half plane zero. If unity gain is not reached sufficiently before the right-half plane zero, then the -180 of phase lag from the LC double pole combined with the -90 of phase lag from the right-half plane zero will result in negating the phase bump of the compensator. The compensator zeros should be placed either before or only slightly after the LC double pole such that their positive phase contributions offset the -180 that occurs
VOUT2
+
ERROR AMP
R3 0.8V FB2 R1 C3
-
VC2 R2 C1
3558 F11
C2
RFB
Figure 11. Error Amplifier with Type III Compensation
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LTC3558 APPLICATIONS INFORMATION
Input Current Limit The input current limit comparator will shut the input PMOS switch off once current exceeds 700mA typical. Before the switch current limit, the average current limit amp (620mA typical) will source current into the feedback pin to drop the output voltage. The input current limit also protects against a short-circuit condition at the VOUT2 pin. Reverse Current Limit The reverse current limit comparator will shut the output PMOS switch off once current returning from the output exceeds 450mA typical. Output Overvoltage Protection If the feedback node were inadvertently shorted to ground, then the output would increase indefinitely with the maximum current that could be sourced from the input supply. The buck-boost regulator protects against this by shutting off the input PMOS if the output voltage exceeds a 5.75V maximum. Buck-Boost Regulator Soft-Start Operation Soft-start is accomplished by gradually increasing the reference voltage over a 500s typical period. A softstart cycle occurs whenever the buck-boost is enabled, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes. This allows seamless output operation when transitioning between Burst Mode operation and PWM mode operation. Buck-Boost Switching Regulator Inductor Selection The buck-boost switching regulator is designed to work with inductors in the range of 1H to 5H. For most applications, a 2.2H inductor will suffice. Larger value inductors reduce ripple current which improves output ripple voltage. Lower value inductors result in higher ripple current and improved transient response time. To maximize efficiency, choose an inductor with a low DC resistance and a DC current rating at least 1.5 times larger than the maximum load current to ensure that the inductor does not saturate during normal operation. If output short-circuit is a possible condition, the inductor current should be rated to handle up to the peak current specified for the buck-boost regulator. The inductor value also affects Burst Mode operation. Lower inductor values will cause Burst Mode switching frequencies to increase. Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but cost more than powdered iron core inductors with similar electrical characteristics. Inductors that are very thin or have a very small volume typically have much higher core and DCR losses and will not give the best efficiency. Table 4 shows some inductors that work well with the buck-boost regulator. These inductors offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information on their entire selection of inductors.
Table 4. Recommended Inductors for the Buck-Boost Switching Regulator.
INDUCTOR TYPE DB3018C D312C DE2812C DE2812C CDRH3D16 SD12 *Typical DCR
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L (H) 2.4 2.2 2 2.7 2.2 2.2
MAX IDC (A) 1.31 1.14 1.4 1.2 1.2 1.8
MAX DCR (m) 80 140 81 87 72 74
SIZE IN mm (L x W x H) 3.8 x 3.8 x 1.4 3.6 x 3.6 x 1.2 3 x 3.2 x 1.2 3 x 3.2 x 1.2 4 x 4 x 1.8 5.2 x 5.2 x 1.2
MANUFACTURER Toko www.toko.com
Sumida www.sumida.com Cooper www.cooperet.com
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LTC3558 APPLICATIONS INFORMATION
Buck-Boost Switching Regulator Input/Output Capacitor Selection Low ESR (equivalent series resistance) ceramic capacitors should be used at both the buck-boost regulator input (PVIN2) and the output (VOUT2). It is recommended that the input be bypassed with a 10F capacitor. The output should be bypassed with at least a 10F capacitor if using Type I compensation and 22F if using Type III compensation. The same selection criteria apply for the buck-boost regulator input and output capacitors as described in the Buck Switching Regulator Input/Output Capacitor Selection section. PCB Layout Considerations In order to deliver maximum charge current under all conditions, it is critical that the backside of the LTC3558 be soldered to the PC board ground. The LTC3558 has dual switching regulators. As with all switching regulators, care must be taken while laying out a PC board and placing components. The input decoupling capacitors, the output capacitor and the inductors must all be placed as close to the pins as possible and on the same side of the board as the LTC3558. All connections must also be made on the same layer. Place a local unbroken ground plane below these components. Avoid routing noisy high frequency lines such as those that connect to switch pins over or parallel to lines that drive high impedance inputs.
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LTC3558 TYPICAL APPLICATIONS
UP TO 500mA USB (4.3V TO 5.5V) OR AC ADAPTER VCC 110k 10F NTC 28.7K 510 100k (NTC) NTH50603NO1 LTC3558 4.7H SW1 CHRG 1.74k PROG SUSP HPWR DIGITAL CONTROL EN1 EN2 MODE GND2 (EXPOSED GND PAD) SWCD2 VOUT2 619k FB2 200k VC2
3558 TA02
BAT PVIN1 PVIN2 10F
+
1 4.7F
SINGLE Li-lon CELL (2.7V TO 4.2V)
1.8V AT 400mA 806k 10pF 10F
FB1 SWAB2 2.2H 3.3V AT 400mA 649k
10F 15k 150pF
Figure 12. Li-Ion to 3.3V at 400mA, 1.8V at 400mA and USB-Compatible Battery Charger
As shown in Figure 12, the LTC3558 can be operated with no battery connected to the BAT pin. A 1 resistor in series with a 4.7F capacitor at the BAT pin ensures battery charger stability. 10F VCC decoupling capacitors are required for proper operation of the DC/DC converters. A three-resistor bias network for NTC sets hot and cold trip points at approximately 55C and 0C. The battery can be charged with up to 950mA of charge current when powered from a 5V wall adaptor, as shown
in Figure 13. CHRG has a LED to provide a user with a visual indication of battery charge status. The buck-boost regulator starts up only after VOUT1 is up to approximately 0.7V. This provides a sequencing function which may be desirable in applications where a microprocessor needs to be powered up before peripherals. A Type III compensation network improves the transient response of the buck-boost switching regulator.
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LTC3558 PACKAGE DESCRIPTION
UD Package 20-Lead Plastic QFN (3mm x 3mm)
(Reference LTC DWG # 05-08-1720 Rev A)
0.70 0.05 3.50 0.05 (4 SIDES)
1.65 0.05
2.10 0.05
PACKAGE OUTLINE 0.20 0.05 0.40 BSC RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED BOTTOM VIEW--EXPOSED PAD PIN 1 NOTCH R = 0.20 TYP OR 0.25 x 45 CHAMFER 19 20 0.40 0.10 1 2 1.65 0.10 (4-SIDES)
3.00 0.10 (4 SIDES) PIN 1 TOP MARK (NOTE 6)
0.75 0.05 R = 0.05 TYP
R = 0.115 TYP
(UD20) QFN 0306 REV A
0.200 REF 0.00 - 0.05 NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
0.20 0.05 0.40 BSC
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC3558 TYPICAL APPLICATIONS
UP TO 950mA 5V WALL ADAPTER 100k VCC 1F NTC 510 100k (NTC) CHRG 887 PROG DIGITAL CONTROL SUSP HPWR MODE EN1 LTC3558 BAT PVIN1 PVIN2 4.7H SW1 324k FB1 SWAB2 2.2H SWCD2 VOUT2 324k 33pF EN2 GND2 (EXPOSED PAD) GND FB2 105k VC2
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+
10F
SINGLE Li-lon CELL (2.7V TO 4.2V)
1.2V AT 400mA 10pF 10F
649k
3.3V AT 400mA 121k
22F
15k 330pF 10pF
Figure 13. Battery Charger Can Charge a Battery with Up to 950mA When Powered From a Wall Adapter
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PART NUMBER DESCRIPTION LTC3550 Dual Input USB/AC Adapter Li-Ion Battery Charger with Adjustable Output 600mA Buck Converter Standalone Linear Li-Ion Battery Charger with Adjustable Output Dual Synchronous Buck Converter Standalone Linear Li-Ion Battery Charger with Dual Synchronous Buck Converter Dual DC/DC Converter with USB Power Manager and Li-Ion Battery Charger COMMENTS Synchronous Buck Converter, Efficiency: 93%, Adjustable Output at 600mA, Charge Current: 950mA Programmable, USB Compatible, Automatic Input Power Detection and Selection Synchronous Buck Converter, Efficiency: >90%, Adjustable Outputs at 800mA and 400mA, Charge Current Programmable Up to 950mA, USB Compatible, 5mm x 3mm DFN-16 Package Synchronous Buck Converter, Efficiency: >90%, Outputs 1.8V at 800mA and 1.575 at 400mA, Charge Current Programmable up to 950mA, USB Compatible Seamless Transition Between Input Power Sources: Li-Ion Battery, USB and 5V Wall Adapter, Two High Efficiency DC/DC Converters: Up to 96%, Full Featured Li-Ion Battery Charger with Accurate USB Current Limiting (500mA/100mA) Pin-Selectable Burst Mode Operation, Hot SwapTM Output for SDIO and Memory Cards, 4mm x 4mm QFN-24 Package
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2-Cell, Multi-Output DC/DC Converter with Seamless Transition Between 2-Cell Battery, USB and AC Wall Adapter Input Power Sources, USB Power Manager Main Output: Fixed 3.3V Output, Core Output: Adjustable from 0.8V to VBATT(MIN), Hot Swap Output for Memory Cards, Power Supply Sequencing: Main and Hot Swap Accurate USB Current Limiting, High Frequency Operation: 1MHz, High Efficiency: Up to 92%, 4mm x 4mm QFN-24 Package USB Charger with Dual Buck Regulators Adjustable, Synchronous Buck Converters, Efficiency >90%, Outputs: Down to 0.8V at 400mA Each, Charge Current Programmable Up to 950mA, USB-Compatible, 3mm x 3mm QFN-16 Package Charges Single-Cell Li-Ion Batteries, Timer Termination + C/10, Thermal Regulation, Buck Output: 0.8V to VBAT, Buck Input VIN: 2.7V to 5.5V, 3mm x 3mm DFN-10 Package
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Hot Swap is a trademark of Linear Technology Corporation.
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32 Linear Technology Corporation
(408) 432-1900
LT 0408 * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
FAX: (408) 434-0507 www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2008

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